Inductor type synchronous motor driving system

ABSTRACT

A system for driving an inductor type synchronous motor for minute control includes an inductor type synchronous motor having first means for driving m numbers of phases and a group of first magnetic teeth, second means equipped with a group of second magnetic teeth; and power feed equipment for feeding power to the driving windings of the individual phases; wherein the minute control characterized by the electrical resolution number R per electrical angle 2m (2m being one fundamental cycle) of the power feed equipment is larger than 3 m points (R 2  3 m) or is stepless; and the power feed equipment feeds electric-waves of current or voltage to the driving windings, the electric-waves being trigonometric functional patterns formed by a control means of the power feed equipment.

BACKGROUND OF THE INVENTION

This invention relates to improvements in inductor type synchronousmotor driving systems for finely controlling the position and therotating angle, in which the vector rotation angle of the supply poweris accurately projected to the mechanical rotating angle and thus highlyprecise resolving control is readily achieved. That is, in the inductortype synchronous motor of nondistortion electromagnetic structure, thevector synthesis theory is strictly established and, at the same time,high accuracy resolution and smooth rotation are realized simply anduniquely through trigonometric functional power feed.

Prior art motors of this type are referred to as the stepping motor,having a plurality of driving windings and a device for feeding themotor rotationally and in a given sequence, with currents whose phasesdiffer from each other. Among these motors is the type having a DC fieldmeans (i.e., DC field winding or permanent magnet), in addition to thedriving windings of respective phases. The inductor type synchronousmotor to be improved by this invention has a first inductor with anarrangement of first magnetic tooth group, and a second inductor with anarrangement of second magnetic tooth group, the motor further havingwinding slots for the driving windings and a unit magnetic path (or amagnetic salient) on the back of the magnetic tooth group of the firstinductor.

In order to increase the number of steps per rotation, prior arttechniques have employed a method of increasing the number of steps, R,(the resolution number R of the vector rotation angle) per cycle(electrical angle 2π) of the feeding current or a method of increasingthe number of teeth, Q₂, of the second magnetic tooth group. Of theformer method, the following methods have been proposed: (1) Method ofincreasing the number of phases, m, of the driving winding; (2) Methodof increasing the number of combinations of windings to which power isfed, that is, n-phase excitation → (n+1)-phase excitation; (3) Method ofincreasing the number of steps, N_(step), where current is fed for therespective phases in stepped waveforms.

These prior art concepts, however, are impracticable for the followingreasons:

The relationship between the torque τ; produced and the rotation angledeviation (load angle) δ in the state of arbitrary power feeding to anm-phase inductor type motor is given as follows according to the torquecurve plotted for the i-phase load angle δ_(i) of torque π_(i) producedon the i-phase during power feed to the i-phase and on the principle ofsuperposition: ##EQU1## where δ_(i) : load angle for i-phase, havingphase difference φ_(i) on i-phase against total reference load angle

i_(i) : feeding current to i-phase

τ_(i) : torque produced on i-phase

The curve of the torque produced on each phase for a given current hasno trigonometric functional characteristic as indicated by curves B andC in FIG. 1. To increase the torque gain (gradient near the origin), thetorque curve has been made as curve B by the use of uniform magneticteeth. These curves have hitherto been considered as trigonometricfunctional characteristic for convenience sake or approximately.Furthermore, the curves offer no linear characteristic with respect tochanges in the current fed.

The torque τ_(i) (δ_(i)) produced at a given load angle δ_(i) withcurrent i_(i) changed follows a nonlinear curve as indicated by curve Ain FIG. 2. This curve shows an example of a characteristic referred toas a reluctance motor, i.e., the so-called variable reluctance type (VRtype) motor having no DC field. This motor has a square characteristicin the small current region, and a saturation characteristic in thelarge current region. The pattern (ratio) itself of the curve varies bythe load angle δ_(i). Therefore, when the current is reduced, thecharacteristic changes from curve B to curve D as in FIG. 1, and theload angle at the maximum torque point shifts by Δ_(p).

In other words, even if the values of currents being fed are changedproportional to each other with respect to the individual phases on thebasis of the state that a given current I₀ is fed to each phase, thestationary balance point moves as indicated by the curve in FIG. 3,resulting in an error θ.sub.ε due to variations in the current value I.This has made it impossible to change the feed current when the motor isdriven under high-resolution control (minute control or verniercontrol). Accordingly, it has also been impossible to reduce the timetaken to increase or decrease motor speed by producing a large torqueduring acceleration or deceleration of the motor.

Furthermore, when current feed per phase is given in n numbers ofstepped waveforms as in FIG. 4(a), the steps Δi_(l) to Δi_(n) differfrom each other and must be adjusted for each motor. FIG. 4(b) shows byexample the period for which the current for one phase is increased; thesolid line i_(n) denotes a current value, and the dotted line Δi_(n)denotes the incremental value of each step. In these patterns, there areno simple functional relationships. In motors with the same number ofphases and similar in construction to each other, their step patternsdiffer from each other. The adjustment of the step patterns for theindividual motor requires extremely intricate skill because onestationary balance point interferes with another stationary balancepoint. Furthermore, it is impossible to change the total feed current Ibecause no proportional relationship is established between the totalcurrent I and each of the steps Δi_(l) to Δi_(n).

For the above reasons, it has been difficult to accurately increase thenumber of resolutions by current ratio. Also, it has been impossible todrive a motor of different capacity or construction by the same powerfeed device or the same power feed control pattern. This has hamperedthe development of standard systems for driving inductor typesynchronous motors.

In prior art techniques, the kind of magnetic teeth which intersect thewinding of one phase is purposely limited to one in order to increasethe torque gain or the maximum torque itself. In high precision minuteresolution control, the feed power is purposely made nontrigonometric infunction. This leads to intricate construction of the power feed deviceand makes its adjustment difficult. This simplifies the constructionresults of large error in minute resolution control, and thesignificance of resolution control is lost.

In the prior art device, there is no linear proportional characteristicbetween the feed current vector rotation angle and the driving force(torque) balance point.

In the device of phase separation type, there is no composite field gapused in common, corresponding to the current vector rotation angle.

The torque (driving force) balance point is not determined byone-dimensional dynamics according to vector synthesis from theelectrical direction of each phase and the produced torque of eachphase.

In the prior art devices, the vector rotation angle is made tocorrespond to the torque balance point θ.sub.τ (although the vectorrotation angle does not essentially correspond to the torque balancepoint in terms of linearity mapping). Such correspondence is attained bynonlogical function means through adjustments. This has requiredintricate adjusting procedures. In addition, power source fluctuation(current variations) affects the adjusted result. Further, suchadjustment is required each time the device is installed in a differentplace. Still further, setting must be changed for each motor used, andthe freedom of the system is considerably limited.

BRIEF SUMMARY OF THE INVENTION

It is an object of the invention to realize minute resolution controlwith simplicity and high accuracy for inductor type synchronous motordevices which perform high-resolution control.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete appreciation of the invention and many of the attendantadvantages thereof will be readily obtained as the same becomes betterunderstood by reference to the following detailed description whenconsidered in connection with the accompanying drawings, wherein:

FIGS. 1 through 4 are characteristic diagrams for illustrating problemsof prior art devices.

FIG. 1 shows one-phase excitation displacement-to-torque characteristic,

FIG. 2 shows a current-to-torque characteristic,

FIG. 3 shows a current-to-stationary point variation characteristic, and

FIG. 4 shows a rotation angle-to-current waveform curve.

FIG. 5 is a rotation vector diagram for illustrating the technicalconcept basis of the invention,

FIG. 6 is a one-phase excitation displacement-to-torque characteristiccurve diagram of a low-distortion inductor type synchronous motor usedfor the purpose of the invention.

FIG. 7 is a current-to-torque characteristic curve diagram of alow-distortion inductor type synchronous motor having a DC field magnetsuitable for the invention,

FIG. 8 is a diagram showing the construction of one embodiment of theinvention,

FIGS. 9, 10, 12 and 14 are distribution pattern and feed power patterndiagrams for illustrating methods of trigonometric functional power feedaccording to the invention,

FIGS. 11, 13, and 15 are circuit diagrams of main circuits for the powerfeed device of the invention,

FIGS. 16 through 20 are enlarged views showing the construction of theinductor and the driving winding slot portion of the low-distortioninductor type synchronous motor, and unit coil flux intersectingrelationship,

FIGS. 21 through 24 are diagrams showing methods of winding the drivingwindings of the low-distortion inductor type synchronous motor used forthe invention.

FIGS. 25(a) to 25(c) are diagrams showing the construction of a DC fieldmagnet of the low-distortion inductor type synchronous motor having a DCfield magnet, suitable for the purpose of the invention,

FIG. 26 is a circuit diagram showing in detail a trigonometricfunctional mode power feed device suitable for the invention, and

FIG. 27 is a diagram for illustrating the operation of the device shownin FIG. 26.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

As for high resolution control of stepping motors, prior art techniqueshave been lacking in fundamental technical concept. In this connection,the outline of the basis on which the technical concept of the inventionlies will be first described.

In inductor type polyphase synchronous motors, the polyphase drivingwinding sets up a driving force by current containing at least apolyphase AC component, to allow polyphase unilateral current orpolyphase bilateral current to pass through. The former type is calledthe single-way type of half-wave type; and the latter, the double-waytype or full-wave type.

Assume that one circulation of current through the driving winding isone cycle, which corresponds to an electrical angle of 2π. Then thedriving winding of each phase should be supplied with a currentcorresponding to its phase. There is an axis in the plane of electricalrotation coordinate corresponding to the phase of the feed current. Inthe three-phase motor, for example, there are three axes, i, j and k, asshown in FIG. 5(a). In this electrical plane, the axial direction of thecurrent fed to the winding of each phase depends upon the constructionand arrangement of the winding of each phase. When, for example, athree-phase motor is driven by two unilateral currents, the fixed axialvalues x_(i) and x_(j) with which the rotation vector x (θ) having anarbitrary rotating angle is obtained signify the current values to begiven to the windings having the corresponding axes. If the compositerotation vector x (θ) plots a circle C whose radius value is 1 (r=1),the values x_(i) and x_(j) projected by the dotted lines parallel withthe mutual axes represent the values of the currents fed to the twocorresponding driving windings. The waveforms of these currents at theangle θ are as shown in FIG. 12(a). If the composite rotation vector x(θ) plots a hexagon C₆ as indicated by the dot-dash line, the waveformsthereof are as shown in FIG. 12(b).

Assume that a three-phase motor is driven by bilateral currents toobtain the desired rotation. In FIG. 5(c), the circle C (r = 2/3) is onewhose radius is two-thirds the given value 1. For the desired rotationvector x(θ), the vector values x x_(i), x_(j) and x_(k) resulting fromvertical projections onto the axes i, j and k from the intersection P ofthe vector x(θ) and the circle C (r = 2/3 ) represent the necessarycurrent values for the respective phases. This three-phase alternatingcurrent is of a waveform which is well-known as shown in FIG. 9(a).While, when the composite rotation vector plots a hexagon C₆, thewaveforms thereof are as shown in FIG. 9(b) or 9(c).

Generally, a multi-axial (or multi-phase) vector can be converted into avector of orthogonal two-axis (d, q) coordinate. This means theobtaining of a composite vector of circle C or n-angled polygon C_(n),as indicated in FIG. 5(d).

Thus, when the axial values x_(i), x_(j), . . . of the composite vectorx (θ) are made to correspond to the currents of the respective phases,the rotation angle θ corresponds to the rotation angle of the stationarybalance point of the synchronous motor on the current vector thereof.When these axial values are made to correspond to the voltages of theindividual phases, the rotation angle θ corresponds to the rotationangle thereof on the voltage or current vector. In this specification,the rotation vector x(θ) which has the foregoing meaning is referred toas a polyphase AC rotation vector for current fed to the polyphasedriving winding, or briefly, rotation vector; and its rotation angle θ,as the vector rotation angle. The term "alternating current" does notmean bipolar current on a particular winding; even if the current forone phase is unidirectional, such current may be given a component ofopposite polarity due to the cause of other phases and can be equivalentto alternating current. The invention places significance on the ACcomponent at least in connection with the above-mentioned alternatingcurrent. This is because a rotation vector is obtainable by the use ofAC component. This AC component will herein be referred to asalternating current (AC) or polyphase AC.

In motor drive by stepping, the vector rotation angle changes in theform of step. The number of steps for one period (one cycle,corresponding to an electrical angle of 2π) of the vector rotation anglewill herein be referred to as the number of resolutions of vectorrotation angle or resolution number R.

In order that the vector rotation angle θ which is defined by theelectromagnetic directivity (magnetic axis) and the current value ofeach phase correspond accurately to the stationary balance point θ.sub.τof the torque produced, the relationship between the torque produced foreach phase and the load angle δ_(i) for each phase must be characterizedby a trigonometric functional curve as shown in FIG. 6. In practice, theexpression of this curve is quite inaccurate although prior arttechniques have been dependent upon such functional curve forconvenience sake, as described by referring to FIGS. 1 to 4. As aresult, the vector rotation angle θ of an electrical quantity(especially current) does not coincide with actual motor rotation angleθ_(m) (especially the stationary balance point).

Most prior art inductor type stepping motors have inductor teethinherent in the individual windings. Substantially, this type of motoris of the phase separation type having independent inductor pairmagnetic paths and gaps (where the yoke or the core back may be used incommon). The motor of this construction has a gap inherent in each phasebut has no common gap (no common space). In other words, the motor hasno common gap for producing a rotating magnetic filed which is a directprojection of the rotation vector x() of feed current as described byreferring to FIG. 5. In such phase separation type motor, there must bea proportional characteristic between the torque _(i) produced for eachphase and the current i_(i). In this respect, the phase separation motorof VR type (FIG. 2) is quite impracticable.

To realize such a proportional characteristic, a method in which a DCfield is provided, and an AC component is supplied separately to thedriving winding is useful, as will be further described later byreferring to FIG. 25. According to this method, the characteristic curvefor τ_(i) and i_(i) can be improved as in curve B in FIG. 7 which is asaturation curve having a proportional linear part C. In addition, thedesired characteristic curve as in FIG. 6 can be obtained by thearrangement that magnetic fluxes of magnetic teeth which differ in thedegree of coincidence are made to interlink with the winding of onephase.

Furthermore, according to the invention, inductor teeth are provided incommon for the driving windings of the individual phases, and gaps areprovided in common, and across the gap, it is possible to produce arotating magnetic field which is a direct projection of the rotationvector x(θ) of feed current (FIG. 5), as will further be described laterby referring to FIGS. 22 and 24. In this case, the proportionalrelationship between the current i_(i) and the torque τ_(i) is notrequired. This is because the direction itself of the rotating fielddepends upon the magnetomotive force itself proportional to the currentfed to each phase. In the phase separation type (as will be describedlater by referring to FIG. 23), the current of each phase is onceprojected on the torque τ_(i), and the rotation angle θ at the compositetorque balance point is determined by synthesizing the torques of theindividual phases, resulting in the problem that nonlinearity isintroduced in the stage where it is projected on the torque τ_(i).However, by producing a rotating field across the common gap, thecurrent (magnetomotive force) vector rotation angle θ becomes directlycoincident with the rotation angle θ at the composite torque balancepoint.

In this manner, the vector synthesizing theory is accurately establishedby the use of the inductor type motor of nondistortion electromagneticconstruction in which magnetic distortion is eliminated. Thus theelectrical vector rotation angle θ is made to accurately reflect thetorque balance point θ.sub.τ and the motor rotation angle θ_(m)accordingly whereby highly accurate resolution is realized throughsimple trigonometric functional power feel control. The invention willbe described in more details in connection with a trigonometricfunctional power feed device, and with an inductor type synchronousmotor of nondistortion electromagnetic construction.

FIG. 8 is a diagram showing an embodiment of the invention applied to aninductor type synchronous motor driving device with the aim to describethe invention in detail. In FIG. 8, the reference Pin denotes a digitalcommand input, 1 a power feed device, 2 a low distortion inductor typesynchronous motor suited for the purpose of the invention, 3 a maincircuit device for the power feed device 1, comprising a power source 5and a solid-state switch 6, and 4 a control circuit device for the powerfeed device 1, capable of generating distribution patterns S forproducing "conducting control signals or reference waveform patterns"which determine polyphase feed currents I or polyphase feed voltage V.The distribution patterns S, as will be described later by referring toFIGS. 9, 10, 12 and 14, are signals for feeding power in half-wave,full-wave or pulse-width modulated wave which collectively representshalf- and full-wave, or in step-approximated wave; and are substantiallyprojection-wise power-amplified into the feed currents I or the feedvoltages V.

Assume that the motor 2 is of three-phase, with bilateral feeding ofthree-phase AC power, and that its currents I or voltages V are of sinewave as in FIG. 9(a). Then the rotation vector locus (or briefly, locus)of the feed power is a circular locus C (r=1) as in FIG. 5(c). When itis a trapezoidal waveform (whose flat section corresponds to anelectrical angle of π/3) as in FIG. 9(b), the rotation vector locus is aregular hexagon (n=6) as in FIG. 5(d). When it is a polygonal-lineapproximation sine wave as in FIG. 9(c) where the electrical angle isπ/3 for each polygonal section with a gradient of ±1, and the electricalangle is π/3 respectively for two polygonal sections each with agradient of ±1/2, the rotation vector locus is a regular hexagon C₆.

(i) Further, these patterns may be of time-ratio modulation waves(pulse-width modulation waves) which represent those shown in FIG. 9.

(ii) Further, these patterns may, in some cases, be of the waveform ofthe distribution patterns S. In other words, these patterns can becontrolled by the main circuit device 3 to agree with the referencepatterns S.

(iii) Further, the patterns can be decomposed into a positive half-waveand a negative half-wave, or into various kinds of components.Accordingly, the distribution pattern S can be decomposed into variouscomponents.

(iv) The most typical example is such that the distribution pattern S isa pulse-width modulation wave representing waveforms shown in FIG. 9 andincludes a conducting signal for positive power feed and a conductingsignal for negative power feed. The component of each pulse-widthmodulation signal consists of components in number corresponding to thenumber of solid-state switches in the power feed device 3. In this case,the pulse-width modulation pattern S is of a conducting control signalfor the solid-state switches, as well as of a pattern for thepulse-width modulation feed voltages V or the pulse-width modulationfeed currents I.

The relationships described in (i) through (iv) above do not depend uponthe number of phases and is irrespective of whether the feed current tothe winding of each phase is unilateral or bilateral. This principle iscommon to operations shown in FIGS. 10, 12 and 14.

FIG. 10 shows patterns for the instance where the motor 2 is oftwo-phase as in FIG. 8; FIG. 10(a) is a sine wave bipolar pattern, FIG.10(b) is a trapezoidal wave bipolar pattern, with the flat section beingan electrical angle of π/2, and FIG. 10(c) is a unipolar trapezoidalwaveform pattern. In FIG. 10, the references V₁ and V₂ denote voltagesof the respective phases, and I₁ and I₂ currents of the respectivephases. The reference I₁ ⁺ indicates the pattern of the first phasepositive winding current, I₁ ⁻ the pattern of the first phase negativewinding (or the third phase) current, I₂ ⁺ the pattern of the secondphase positive winding current, and I₂ ⁻ the pattern of the secondnegative winding (or the fourth phase) current. The waveform of FIG.10(a) corresponds to the two-axis (d,q) coordinate of FIG. 5(d) andexhibits a circular locus C, and those of FIGS. 10(b) and 10(c) exhibitan octagonal locus C₈.

An example of a power feed device for this motor is illustrated in FIG.11, in which the references 5₁ and 5₂ denote DC power sources, 6_(1a)through 6_(2b) solid-state switches, and 1₁ and 1₂ or 1_(1a) through1_(2b) windings of individual phases.

In FIG. 11(b), the reference 9 denotes a solid-state circuit similar tothe one shown in FIG. 13 which has a solid-state switch 7 and a diode 8additionally for each arm.

FIG. 12 shows patterns for the instance where the motor 2 is ofthree-phase and the power feed device is of three-phase half-wave feed.These patterns are of three-phase current (each current i₁, i₂ and i₃).

The pattern of FIG. 12(a) exhibits a circular locus C of FIG. 5(b), eachphase pattern consisting of an envelope of two half sine-waves, the twohalf-sine-waves for one phase having a phase difference by an electricalangle of π/3 from one another.

The pattern of FIG. 12(b) exhibits a regular hexagonal locus C₆ of FIG.5(b), of which the waveform for one phase is a trapezoidal waveformcomprising a flat section corresponding to an electrical angle of 2/3πand a slope of 1/3π.

The pattern of FIG. 12(c) is one obtained by polygonal lineapproximation of the pattern of FIG. 12(a) and coomprises a section witha gradient of ±1 and a section with a gradient of ±1/2. In the middle,the pattern has a return point P₁ or an intersection P₂ at which twophases intersect with each other. One cycle (2π) of it is divided into12 parts. This pattern exhibits a regular dodecagonal locus C₁₂.

FIG. 13 shows an example of connection of the main circuit for thecurrent feed device as shown in FIG. 12. In FIG. 13, the reference 5₁denotes a main source, and 5₂ ^(') an auxiliary source allowing the flowof current for the device 5₁ through, for example, a voltage converter.For this voltage converter, a chopper may be used. The references 8₁through 8₃ denote rectifier elements, 7₁ through 7₃ solid-state switchessuch as transistors and thyristors, and 1₁ through 1₃ driving windingsof the respective phases.

A simple power feed device of a three-phase single way has beendescribed by referring to FIGS. 12 and 13. This device is capable ofoffering a circular locus C as in FIG. 12(a) and a dodecagonal locus C₁₂as in FIG. 12(c) with a good cost-performance factor. The one offering ahexagonal locus C₆ as in FIG. 12(b) makes vector rotation resolutionavailable equivalent to those available as in FIGS 9(b) and 9(c), with agood cost-performance factor.

FIG. 14 shows an example of how two-phase (angle difference 120°, 60°)power is fed to a three-phase motor. FIG. 14(a) to 14(c) are patternswith one phase removed from those of the three-phase as in FIGS. 9(a) to9(c). FIG. 15(a) shows an example of a main circuit used for power feedwith these patterns. In FIG. 15(a), the references 6₁ and 6₂ denotesolid-state switches (bilateral switching modules) connected in thecircuit similar to the switch circuit as in FIG. 11(a). FIG. 15(b) showsconnections (i), (ii) and (iii) for motor windings corresponding tothese switches; that is, three-phase Y connection, three-phase Δconnection, and three-phase V connection. The patterns as in FIG. 14,together with the connection arrangement as in FIG. 15, offer rotationvector loci (such as circular locus C and regular hexagonal locus C₆)equivalent to those shown in FIG. 9. This embodiment serves to simplifythe power feed device and is suited for smaller outputs.

For operation where the same type of three-phase motor is driven underthe patterns of FIG. 9 (for example, by a three-phase bridge inverter)at a speed lower than its maximum speed, the patterns and arrangement asin FIGS. 14 and 15 may suitably be utilized. Under normal operation, thepatterns of FIG. 9 are used with a Y connection; while, under operationat a torque lower than the maximum torque, the patterns of FIG. 14 areused with a Δ connection as in FIG. 15.

As described above, the locus of rotation vector x(θ) can be drawn intoa circle or n-angle polygon (n=6,8,12,...) when the voltage V orcurrents I of the foregoing patterns, or the voltage or current ofpulse-width modulation waveforms representing the foregoing patterns issupplied to the motor windings of the individual phases. Furthermore, itbecomes possible to produce a distribution pattern S (reference waveformor turn-on signal for the solid-state switch) for voltage or current tobe supplied. The distribution pattern S, i.e., the vector rotation angleθ of the power feed voltage or current, can be made to correspond to thedigital input Pin. Generally this is achieved by the use of arecurrently operable counter, a trigonometric function generator, a sinewave oscillator, a signal generator at a plurality of frequencies, atleast one of which is variable, or a like device. Another concreteexample is one in which ring counters are used and their outputs aresuitably synthesized whereby a pattern approximate to a polyphase sinewave is obtained. When an n-nary notation ring counter is used to countits input up or down by a pulse train of two different frequencies, itbecomes possible to obtain a pulse-width modulation pattern S whichrepresents the rotation vector x(θ) of an n-angle polygon. By filteringthis pattern S, a variety of segment-approximated patterns can beobtained. Furthermore, when the solid-state switches used are of Nnumbers being equal to the number n (N=n), these switches can be on-offcontrolled directly by the output of the n-nary ring counter. Thisapproach is most simple and effective. In this manner, the power feeddevice 1, the main circuit device 3 and the distribution patterngenerator 4 can be realized.

FIG. 26(a) shows a concrete example of the power feed device 1corresponding to the embodiment shown in FIG. 8. In FIG. 26(a), thevoltage of a DC power source 300 is on-off controlled by chopper 301aand 301b. The DC voltage Vdc is controlled by flywheel rectifierelements 303a and 303b. The resultant voltage is smoothed by DC reactors304a and 304b and a capacitor 309. The DC current Idc and DC voltage Vdcare detected by detectors 306 and 307 respectively and then comparedwith the speed interlocking voltage commmand Vs and the signal Is whichsets a current setter 308 or with the adaptation command signal Is'wherein the ratio of turn-on time of the chopper 301a to that of thechopper 301b is controlled by a chopper control means 302.

This power feed device comprises a group of solidstate switches 311 to316 for the purpose of distributing DC power supply voltage to theindividual phases of the inductor type synchronous motor 2, and arectifier 305 for clipping the solid-state switch voltage and forfeeding back the motor reactive current. The choppers 301(a) and 301band the solid-state switches 311 to 316 may be of transistors orthyristors.

The power feed device 1 further comprises a gate amplifier or a basedriving amplifier 320 for transmitting to a preamplifier, underisolation, signals S₁ to S₆ which are to turn on and off the solid-stateswitches 311 to 316.

The turn-on signals S₁ through S₆ are of the distribution pattern S ofFIG. 8 and can be time-ratio-modulated to modulate in time-ratio theturn-on time of each of the solid-state switches 311 to 316.

The distribution pattern S (S₁ to S₆) which serves as turn-on signals isavailable from the distribution pattern generator 4. The distributionpattern generator 4 comprises a reversible ring counter 405 such as ashift register, up-down counter, decoder, AND-OR logic element forconverting the signal of the state of the ring counter, into a suitablepulse width (the ratio of output time to one cycle) in a suitablerecurring order, frequency dividers 404a and 404b, pulse train adders(OR elements) 403a and 403b, a carrier pulse generator 401, and acarrier pulse gate element (AND element) 402. In FIG. 26(a), the numeral500 denotes a command means.

A positive rotation command pulse PinF comes in from the terminal F, anegative rotation command pulse PinR from the terminal R, and a finecontrol mode command pulse FINE from the terminal C. When the pulse FINEcomes in, a carrier pulse Pce passes through the gate 402 and goes tothe pulse train adders 403a and 403b. The frequency of the carrier pulsetrain Pce is D times as high as the time-ratio modulation frequencyf_(PWM). The frequency dividers 404a and 404b divide a given frequencyat a rate of 1/D.

Assume a low speed region where f_(inf) <<f_(ce) at the input ofpositive rotation command PinF. Only the carrier pulse Pce is applied tothe pulse adder 403b, and the frequency of output pulse P₂ of thefrequency divider 404b is: f₂ = f_(ce) /D = f_(PWM). This frequency isshown in FIG. 27(b), with period T. The input pulse train PinF is shownin FIG. 27(a). The pulse train adder 403a receives the pulse train PinFand the carrier Pce, and the summed pulse train is supplied to thefrequency divider 404a. The frequency f₁ of the output pulse train P₁ ofthe frequency divider 404a is: f₁ = (f_(inf) + f_(ce))/D = (f_(PWM) +f_(inf) /D), which is shown in FIG. 27 (c). Thus, with the pulse trainP₂ taken as a reference, the time position (pulse phase) of the pulsetrain P₁ advances by ΔT each time the command pulse PinF comes in, whereΔT = T/D. The ring counter 405 is an N-nary reversible counter (where N= 6, in this example). When the input is only the pulse train P₁, thering counter 405 generates the H and L signal at the output terminal inphase sequence. When the output cycle is 2, the H-time T_(H) --π, andthe L-time t_(L) --π, in this example. For instance, with the input of asingle count-up pulse train P₁, the first output S₁ is a positivehalf-wave signal indicated by the solid line, and the fourth output S₄is a negative half-wave signal indicated by the solid line as in FIG.27(f).

Assume that the pulse trains P₁ and P₂ are applied to the count-up inputCU and the count-down input CD, respectively, of the ring counter 405,and that the simultaneous pulses indicated by Δ are eliminated as inFIG. 27 (b) and (c). Then the ring counter output S₁ gives a positivehalf-wave signal as in FIG. 27(d), and the output S₄ gives a negativehalf-wave signal.

The solid-state switches 311 and 314 are turned on by the outputs S₁ andS₄ respectively. The solid-state switches 311 to 316 are turned on bythe patterns S₁ to S₆ being the same as the signal S₁ in waveform(positive half-wave) and having a phase lag by 1/6 cycle behind oneanother. As a result, the feed potential EA at the output terminal A ofthe phase A assumes a waveform as in FIG. 27(d). The mean potentialwaveform at the terminal A is indicated by the dotted line in FIG.27(d). At other output terminals B and C there occurs a three-phaseoutput potential in a waveform similar to one shown in FIG. 27(d) with aphase lag by 4/3π behind one another. Accordingly, the output linevoltage V_(AB) becomes as shown in FIG. 27(e). The mean value thereof isin a trapezoidal waveform indicated by the dotted line in FIG. 27(e),which is of the pattern shown in FIG. 9(b). The motor current issmoothed by the motor inductance; in the case of a Δ connection motor,the phase current comprises a small amount of pulsating component,centering the trapezoidal waveform indicated by the dotted line in FIG.27(e). While, in the case of a Y connection motor, the phase currentcomprises a small amount of pulsating component, centering the polygonalline waveform as in FIG. 9(c), which is the same as the waveform ofoutput line i_(A) in the example shown in FIG. 26(a).

In the low speed region, the motor and circuit resistances serve as animportant factor and hence the DC feed current I_(dc) determines thewave heights of AC output currents i_(A), i_(B) and i_(C). In the highspeed region, the AC output current is detected and its value can becontrolled as in the case with I_(dc). In other methods, the AC feedcurrent value I_(ac) is detected and can be controlled.

The embodiment as shown in FIG. 26(a) may be modified so that the ACfeed currents i_(A) to i_(C) are time-ratio-modulated. In such case, thecurrent i_(A) assumes the waveform as in FIG. 27(e). For thismodification, the capacitor 309 is removed, the DC terminal of therectifier 305 is connected in parallel to the DC power source 300, andthe output time (conduction time of the solid-state switch) is set to2/3 π. This modified embodiment is illustrated in FIG. 26(b).

In FIG. 27, when the command pulse Pin stops at the timing indicated bythe "stop" arrow, the pulse trains P₁ and P₂ afterward become only apulse of a frequency division of the carrier pulse Pce and stand at thesame frequency f₁ = f₂ = f_(PWM) = f_(ce) /D. Therefore the phasedifference between P₁ and P₂ remains in the past state. Under thisstate, the ring counter 405 does not advance, only repeating up anddown. The distribution patterns S₁ and S₄ become fixed time-ratiosignals as indicated in FIG. 27 after the timing "stop" of arrow mark.Thus the ratio of the feed current i_(A) is maintained. Consequently,the motor stops at the rotating field rotation angle θ or the drivingtorque balance point θ_(m).

In FIG. 26, one cycle of electrical angle 2π is resolved at a ratio of1/N by the ring counter 405. This is further resolved at 1/D bysuperposing the outputs of frequency dividers 404a and 404b and thecarrier pulse train Pce on each other. This is because the turn-on timeratio is controlled on the basis of ΔT/T = 1/D. In other words, thenumber of resolutions R of an electrical angle 2π is a whole: R (1/D·N).

For these reasons, the electrical angle can be resolved at R = 20 to 600without limitation on the number of phases, such as in the case of athree-phase motor.

When the number of rotor teeth of a motor is Q₂, one rotation can beresolved at a ratio of 1/D·N·Q₂ or of 1/1/2D·N·Q₂. The value of Q₂ mayrange from 10 to 200. (In larger motors, the value of Q₂ can be larger).

If the inductor type synchronous motor is driven at a high speed as inthe universal motor, the output waveform will become high frequency.Therefore it is inefficient for the time-ratio modulation pattern as inFIG. 27(d) and (e) to be maintained by increasing the carrier frequencyproportionally (f_(ce) ∝ f_(in)) in the high speed region. In otherwords, load or loss increases on the side of solid-state switches.

According to the invention, two methods are considered to solve theproblem. One method is such that the carrier frequency is fixed ornearly fixed at a relatively low frequency. In this case, if f_(in) >>f_(ce) in the high speed region, it becomes possible to obtainsquare-wave distribution patterns S₁ (H) and S₄ (H) as in FIG. 27(f). Atthe same time, the potential EA(H) at the output terminal assumes thewaveform (f) in FIG. 27. As a result, the line output voltage V_(AB) (H)assumes the wave-form indicated by the dotted line in FIG. 27(g), andthe output line current i_(A) assumes the polygonal line approximationsine wave indicated by the solid line in FIG. 9(c).

The other method is such that the carrier frequency f_(ce) is eliminatedat a speed higher than a given value. The carrier frequency may beconstant as in the first method. The carrier frequency is chosen to bef_(ce) >> f_(res) as in the first example, where f_(res) indicates thenatural frequency (resonance frequency) of the synchronous motor.

When the frequency f_(in) of the input command pulse PinF or PinR ishigher than the resonance frequency f_(res) (f_(in) >> f_(res)), themotor 2 rotates smoothly and minute control is not needed. Therefore,when the command indicates a speed higher than a given value, the FINEsignal is released and the carrier pulse train Pce is shut out by thegate 402.

In the above manner, the motor operates as a square-wave inverter wherethe distribution patterns S₁, S₄, the potential E_(A) at the outputterminal, the line output voltage V_(AB) (H), and the output linecurrent i_(A) (H) are as shown in FIG. 27 (f) and (g). Thus, byreleasing minute control at a speed higher than a given value, the motorcan be driven at a higher speed. Under the same maximum speed condition,minute control is available at an extremely low speed, permitting themotor to be driven over a wide range of speed.

In the foregoing embodiment, the distribution pattern S istime-ratio-modulated and corresponds directly to the turn-on controlsignal, i.e., the feed voltage or feed current. This enables the deviceto be considerably simplified.

Generally there are available a variety of distribution patterngenerators, among which the desirable one comprises a frequency signalgenerator 410 capable of generating two frequencies f₁ and f₂. Thesefrequency signals contain a pulse component or at least an AC component.These signals can be converted into a distribution pattern S by a knownsimple means.

More specifically, one aspect of this approach is such that the firstand second frequency signals f₁ and f₂ can contain frequency informationand relative frequency information, which are projected on the frequency(fundamental frequency or carrier frequency) of the distribution patternS and hence on the feed power AC frequency.

The second aspect is such that it contains relative phase (phasedifference) information, which is an integral value or an analog valueof the relative frequency (at least f₁ - af₂ where a is a proportionalconstant). The relative phase is of a generalized positional dimension,whereas the frequency is of a generalized speed dimension. Hence therelative frequency is a value of dimension permitting projection into anelectrical angle, a rotating field rotation angle or a balance point(stationary position) of the driving force of the synchronous motor 2.

Accordingly, when at least two frequency signal generators are provided,and the frequency of one of the two is made variable, the signal canreadily be converted into various distribution patterns S suited forminute control for the purpose of the invention. The embodiment shown inFIG. 26 is one example showing the above principle. Another concreteexample is such that a distribution pattern having a relative frequency(f₁ - f₂) and a minute waveform pattern (minute relative phaseinformation) can be obtained by amplitude modulation between twosine-wave frequency signals or by heterodyne modulation. Also, thesignal can be converted into a distribution pattern having a relativefrequency (f₁ - f₂) and a minute waveform pattern (minute relative phaseinformation) by on-off modulating one sine-wave frequency signal bysquare-wave frequency signal (where the on-off modulation corresponds tomultiplication in a synchronous rectifier or analog switch circuit).These approaches are simple yet desirable for the purpose of generatingvarious patterns as shown in FIGS. 9, 10, 12 and 14. The foregoingfrequency signals can readily be generated by a variety of known pulsegenerators or oscillators, analog frequency converters, or the like.

A continuous curvature pattern waveform, a polygonal line waveform or aminute step waveform can be converted into a distribution pattern(turn-on control signal) which has been time-ratio-modulated by atime-ratio modulation means. The distribution pattern is substantiallypower-amplified and projected on the electrical quantity of the feedpower.

One embodiment of the invention applied to a power feed device 1, andits effective approaches have been described in detail by referring toFIG. 8.

To meet the function of the power feed device 1 as in FIG. 8, theinductor type synchronous motor 2 must accurately follow the rotationangle of the AC component vector x(θ), and minute controls thereof mustbe accurately reflected and projected to the driving force (torque)balance point θ.sub.τ. Otherwise, the performance quality of theforegoing power feed device cannot be fully exhibited. Various factorsand improvements for the inductor type synchronous motor needed forminute and accurate driving control according to the invention will bedescribed in detail, together with novel effects available when themotor is operated in combination with the power feed device 1.

FIG. 16 shows one embodiment of the invention applied to an inductortype synchronous motor for which magnetic field modulation is effectedby the use of an inductor. This corresponds to a linearizedcross-sectional view of part of a linear motor or of part of a disk typemotor. In FIG. 16, the reference 100 denotes a first inductor, and 200 asecond inductor. The second inductor comprises a core back 220 and aplurality of second magnetic tooth group 210 projected on the surface ofthe core back 220. Assume that the pitch of the second magnetic teeth isλ₂. In the disk type structure, it may be so arranged that the secondcomprises magnetic segment group (which corresponds to the magnetictooth group) implanted on a support body (support base) without usingthe core back 220, and the support body is interposed between the firstinductor 100.

The first inductor 100 has a first magnetic tooth group 110 with pitchλ₁. On the back of the first inductor are unit magnetic path (magneticshunt) group 120 to 120n, and slot group 130 to 130n for the windings.The first group of magnetic teeth are projected opposite to the gap ofthe unit magnetic path. The unit magnetic paths 120 to 120n are linkedto each other by a core back 150. It may be so arranged that the unitmagnetic paths forming pairs of N-S poles are linked, pair by pair, by acore back magnetic path (not shown). This construction is suitable whenU-shaped unit-magnetic paths are disposed or C-shaped unit-magneticpaths are disposed across the disk type second inductor.

When the pitch of the first magnetic tooth between the unit magneticpaths 121 and 122 which are separated from each other by the windingslot 132 is λ₃. Then λ₃ = k₁ λ₁ (where k₁ = 1, 2, 3, ...). When k₁ = 1,this value is the removal number of the first magnetic tooth, i.e., theremoval number per unit magnetic path interval. FIG. 16 shows an examplewhere k₁ = 1 and hence the removal number is 0. The unit magnetic pathinterval can be matched with the pitch λ₂ of the second magnetic toothby the arrangement where λ₃ = (λ₁ + k₂ λ₂) where k₂ = 0, 1,2, ... Thisprinciple is applicable to other unit magnetic paths. For instance, whenλ₃ = (λ₁ + k₂ λ₂), it becomes possible to determine the unit magneticpath interval individually depending on conditions.

Conductor wires for the driving windings are inserted in the slots 131to 131n respectively, thus forming a group of unit coils 140 with asuitable coil pitch. The first pitch unit coil 141a is wound on the unitmagnetic path 122. The second pitch unit coil 142a or 142b is wound ontwo adjacent unit magnetic paths 121 and 122 (or 122 and 123), andintersects the magnetic fluxes of the two unit magnetic paths. Thesecond pitch unit coil 142b is equivalent to the series of the firstpitch unit coils 141a and 141b which are wound on the unit magneticpaths 122 and 123 respectively. The third pitch unit coil 143 is woundon three unit magnetic paths 121, 122 and 123 and intersects themagnetic fluxes of these unit magnetic paths. Generally, the k₃ -thpitch unit coil which intersects the fluxes of k₃ numbers of unitmagnetic paths can be formed.

The driving windings 1_(l) to 1_(m) for one phase are formed byconnecting a single or plural unit coils 140 serially or in parallel.

When λ₃ = λ₁ in the embodiment in FIG. 16, a coincidence or adiscoincidence occurs between the first and second magnetic teeth ateach λ₁₂ (not shown) which is the least common multiple of the firstmagnetic tooth pitch λ₁ and the second magnetic tooth pitch λ₂. The samedegree of coincidence occurs between other first and second magneticteeth, also at the pitch λ₁₂. In other words, the degree of coincidence(i.e., the permeance of one of the first magnetic teeth to the secondinductor) differs with respect to the spatial position X (or spatialangle θ') and changes recurrently (periodically). The recurrent pitch isλ₁₂.

The degree of coincidence differs on the first magnetic teeth withrespect to one another within one cycle λ₁₂. One degree of coincidence(permeance) can be differentiated from another by the following value(absolute value) and directivity (polarity); first, the absolute valuea_(s) of an area of one of the first magnetic teeth 110 opposite to oneof the second magnetic teeth 210, and second, the directivity causingthe second inductor 200 to be moved in a given direction (positivedirection). Thus the difference in the degree of coincidence can bedetermined by the polarity of the differentiated value (varying ratio)de_(s) /dx (where dx is a positive direction) according to whether theabsolute value a_(s) is on the increase or decrease with the movement ofthe second inductor 200 in a given direction. Accordingly, thedifference between the degrees of coincidence can be found by therelative positional relationship, and hence the kind of the degree ofcoincidence can be defined according to the differentiated result. Thekind of the degree of coincidence will be described below. The degree ofcoincidence on one of the first magnetic teeth (reference magneticteeth) changes periodically with the relative motional displacementθ_(m) of the first inductor 100 to the second inductor 200. When thevalue of periodic functional change is f(θ_(m)), the degree ofcoincidence a_(si) on the other arbitrary magnetic tooth is given by thefollowing equation. Assume that the flux of the i-th of one of the firstmagnetic teeth is φ₁.

    a.sub.s1 = f(θ.sub.m - ζ.sub.11) ∝ φ.sub.1

    a.sub.s2 = f(θ.sub.m - ζ.sub.12) ∝ φ.sub.2 (2)

    a.sub.si = f(θ.sub.m - ζ.sub.1i) ∝ φ.sub.i

where ζ_(1i) is the spatial periodic phase difference between thereference first magnetic tooth and the i-th one of the first magnetictooth.

In other words, the flux of each magnetic tooth is magneticallymodulated by the degree of coincidence a_(si). The mean fluxdistribution across the gap assumes a pattern which is magneticallymodulated by the distribution of the degree of coincidence a_(si).

The kind of group of first magnetic teeth 110 as in FIG. 16 isdetermined by the foregoing spatial phases ζ₁₁ to ζ_(1i). The number ofkinds H thereof is:

    H = λ.sub.12 /(λ.sub.1 - λ.sub.2) (3)

One of the first magnetic teeth 111 is nearly the same as another 112 inFIG. 16 with respect to the absolute value of the degree of coincidence,but differs from the tooth 112 as to directivity (polarity) and phase φin Eq. (2).

The first pitch unit coil 141a, for instance, intersects the fluxes of 8different ones of the first magnetic teeth. The third pitch unit coil143 covers 12 different ones of the first magnetic teeth. When theseries of the first pitch coils 141a and 141b is used for one phase ofthe driving winding, this winding intersects the fluxes of 8 differentones of the first magnetic teeth. While, when the series of the secondpitch coils 142a and 142b is used for one phase of driving winding, thiswinding intersects the fluxes of 12 different ones of the first magneticteeth. The flux intersecting ratio (the number of flux intersections) islarger by four of the first magnetic teeth installed on the unitmagnetic path 122 than by eight of the first magnetic teeth installed onthe unit magnetic paths 121 and 123. This is equivalent to an increasein the area of a magnetic tooth facing the gap, the flux intersectingratio of the magnetic tooth being large.

In the embodiment as in FIG. 16, as described, fluxes of many kinds ofmagnetic teeth affect (intersect) each other, with the result that therelationship between the deviation angle θ_(i) regarding the i-th phaseand the torque τ_(i) for the i-th phase is markedly improved as in FIG.6 as opposed to the one shown in FIG. 1 wherein the magnetic teeth ofone kind intersect fluxes. The reason for this is as follows.

As indicated by Eq. (2), h numbers (h = H) of fluxes of magnetic teeth,i.e., degrees of coincidence on different phases, are compositelyintersected with the driving winding of one phase. When the number ofintersections of the j-th phase driving winding with the i-th magnetictooth φ_(i) is W_(i), the number Ψ_(j) of intersections of the j-thdriving winding with fluxes is: ##EQU2## where i = 1 to h-th magnetictooth intersects the j-th phase. While, when the periodic function isf(θ_(m)), then ##EQU3## where φ₀ : mean tooth flux (DC component)a.sub.ν : ratio for DC component of the ν-th order component

Assume that the fluxes intersecting the j-th phase are φ₁ to φ_(h). Thenthe phase deviations ζ₁₁ to ζ_(1h) lie in the range of ζ_(j) ± Δζ, whereζ_(j) is the deviation from the reference phase of the j-th phasedriving winding. Assume Δζ < π/2. Accordingly, for Ψ_(j), the DCcomponent and the first order component are emphasized by synthesis ofthe order ν and different ζ_(1i) and thus the higher order component iscancelled. It is effective that the center magnetic tooth is intersectedwith a larger number of turns for W_(i). (In FIG. 16, the coils 142a and142b are connected in series, or the coils 141a and 143 are connected inseries to each other.) As a consequence Eq. (4) may be expressed asfollows, with no appreciable error.

    Ψ.sub.j = Ψ.sub.0 {1 - A.sub.1 cos (θ.sub.m - ζ.sub.j)} (6)

where ##EQU4## the number of total flux intersections of fixed portion(mean flux ratio value) ##EQU5## content ratio after synthesis of firstorder component (constant)

    π/2 > Δζ > (ζ.sub.1i - ζ.sub.j) > - Δζ > - π/2

The differential value Ψ = d Ψ_(j) /dζ_(m) is:

    Ψ.sub.j = Ψ.sub.0 A.sub.1 Sin(θ.sub.m - ζ.sub.j) (7)

It is apparent that the torque curve as in FIG. 6 can be obtained bysubstituting the i-th one with the j-th one and by determining δ_(i) =(θ_(m) - ζ_(i)), because the torque is proportional to Ψ_(j). Equation(7) signifies that the internal velocity e.m.f. due to the relativemotion dθ_(m) /dt becomes a sine wave. In other words, the nondistortioninductor type synchronous motor of the invention has a sine-waveinternal c.m.f. as in those of general power use.

When the following current i_(j) of the j-th phase is passed through,

    i.sub.j = I.sub.1 Sin [θ - ζ.sub.j - (π/2)] (8)

then the following equation is obtained since the j-th phase torque isproportional to the product of Ω_(j) and i_(j).

    τ.sub.j = Ψ.sub.0 A.sub.1 I.sub.1 1/2{(Sin (θ - θ.sub.m) - Sin(θ + θ.sub.m -2 ζ.sub.j)} (9)

Generally, in the case of m-phase, the following is chosen: ##EQU6##Thus the total torque τ is led from Eqs. (9) and (10): ##EQU7##Accordingly, the second term in the braces of Eq. (9) is cancelled,where (θ - θ_(m)) represents the total load angle δ. The motor rotationposition θ_(m) at which the torque τ is zero is the torque balance point(electromagnetic force driving force balance point) θ.sub.τ. ##EQU8##where θ_(m) satisfying τ = 0 As in Eq. (8), the angle θ is the vectortorque rotation angle (FIG. 5) of polyphase AC, and the electromagneticforce balance point fully corresponds to the vector rotation angle ofpolyphase AC.

One prior art problem lies in the presence of a large error (distortion)which cannot be approximated by Eqs. (6) and (7).

The reason for this is because the prior art construction has magneticteeth of one single kind, exhibiting a τ_(i) - δ_(i) characteristiccurve as in FIG. 1, without having the function of cancelling the higherorder component.

According to the invention, the operation of an inductor typesynchronous motor is made to correspond to the vector rotation angle θof the feed AC only when many kinds of magnetic teeth have a fluxintersecting relationship. In this concept, the vector composite theoryis exactly established. It is important that many kinds of magneticteeth be intersected with fluxes to permit the motor to be rotateduniquely and accurately in response to the electrical quantity vectorrotation angle θ.

Whereas, in prior art techniques, it is necessary that the currents ofthe individual phases be changed non-trigonometrically in function orthe currents be minutely adjusted as the stationary point is checked allthe time, in order to achieve accurate and minute rotation of the motor.Furthermore, because the vector rotation angle θ of the feed current isnot reflected directly on the stationary balance point θ.sub.τ, theresolution accuracy of the stationary balance pointis lowered if thecurrent representing value (wave height value) I₁ as in Eqs. (8) and(11) is changed.

According to the invention, the current value I₁ serves as theproportional coefficient of torque, causing no deviation of the vectorrotation angle, i.e., the stationary balance point. This readily permitsthe current value I₁ to be suitably varied and controlled.

FIG. 17 shows another embodiment of the invention wherein the firstmagnetic tooth pitch λ₁ and the second magnetic tooth pitch λ₂ are bothλ, within one unit magnetic path. The distance S₂ between unit magneticpaths differs from the nonmagnetic width S₁ of the first magnetic tooth.Here the pitch λ₃ is not a multiple of the integer of the pitch λ₁. Inthis sense, the degree of coincidence is deviated for unit magneticpaths different from one another. In this case, the number H of kinds ofdegrees of coincidence is reduced. The unit coil is wound on a pluralityof unit magnetic paths. In this arrangement, the number h of the kindsof magnetic teeth which intersect the driving winding of one phaseincreases and hence it is desirable that a certain numbers of unit coilsbe connected in series to each other so that they intersect the fluxesof h-number of unit magnetic paths; for instance, the coils 142a and142b, or 143a and 143b, or 142a and 142b and 142c, or 143a and 143b and143c are connected serially.

FIG. 18 shows a method for increasing the number of kinds of degrees ofcoincidence for one unit magnetic path. The tooth widths t₁, t₂ and t₃may be differentiated from each other, or the tooth centers may bedifferentiated, in addition to different tooth widths, such as teeth 110and 110₃ which are equal to each other in respect to tooth width butdiffer from each other in respect to the tooth center and phase ζ_(1i).The arrangement concerning the number of kinds of degrees of coincidenceis described in Japan Patent Application No. 48-69474 (1973).

FIG. 19 shows another method for increasing the number H of the kinds ofdegrees of coincidence, wherein FIG. 19(a) is a side view of the secondinductor 200, with the relative motion direction X(θ') taken as thelongitudinal direction. FIG. 19(b) is a view taken of the facing gapswith their surfaces up, wherein the reference L denotes the length of acylindrical second inductor in the axial direction, which, for instance,corresponds to the laminated thickness. The reference Δζ' denotes thevalue of deviation by Δζ' in the relative motion direction of the toothwithin he length L. This deviation is within the value λ/2 (1/2pitch).By this arrangement, the number of kinds of degrees of coincidence canbe made equal to the number of plates of the laminated cores.Substantially, the number of kinds of magnetic teeth can be arbitrarilyincreased. Instead of continuously deviating them, they may be deviatedin steps. These arrangements are described in Japan Patent ApplicationNo. 48-72603 (1973).

The above methods of increasing the number of kinds of first magneticteeth can be effectively applied to the embodiments shown in FIGS. 16and 17.

FIG. 20 is a partial view of a winding method wherein the nonmagneticspace between the first magnetic teeth is expanded where conductors areinstalled. By this arrangement, the number of unit coils can beincreased, the value W_(i) Eq. (4) can be changed trigonometrically infunctional (in the form of step approximation) according to the i-thnumber, and thus the effect of cancelling the higher order componentagainst the fundamental component can be enhanced. In other words, thefundamental component can be increased when the same cancelling effectis considered. Furthermore, characteristics can be improved by reducingthe leakage inductance.

Described above is the principle of the invention wherein the number ofkinds of magnetic teeth (kinds of degree of coincidence) which intersectthe fluxes of the driving winding of one phase is increased for thepurposes of the invention.

Next, the effect of the invention in connection with the arrangementthat the magnetic teeth intersect a plurality of unit magnetic paths tomake it possible to increase the coil pitch for one phase will bedescribed below, together with the concept of the overall windingconstruction in a cylindrical motor.

FIG. 21 is a circular pattern showing a method of using a short-pitchwinding having no DC field. In FIG. 21, the reference 130 denotes awinding slot, and the arrow indicates the region where the degree ofcoincidence between the first magnetic tooth 110 and the second magnetictooth 210 is large. Detailed portions of the magnetic tooth areindicated roughly by the dotted lines; in practice, there are adistribution of magnetic teeth as shown by an enlarged diagram indicatedin the circuit of dot-dash lines. (This portion will be shown briefly inthe succeeding embodiments.)

FIG. 21 shows an example of three-phase 6-slot 2-pole construction,wherein unit coils a and a are windings of A-phase and form a pair ofmagnetic poles, N and S (or S and N when the current is inverse), forthe second inductor 200. Similarly, unit coils b and b are windings ofB-phase, and unit coils c and c are windings of C-phase. There may bethe combination of unit coils indicated in the parentheses. Thisarrangement, however, causes an eccentric magnetomotive force. Hencethis is desirable for applications where there are more than twice thenumber of poles or the number of slots.

FIG. 22 (a) and (b) show methods of winding in the form of long-pitchunit coil; (a) is 120° pitch for intersecting two unit magnetic paths,and (b) is 180° pitch for intersercting three unit magnetic paths. InFIG. 22, the polarities of the winding conductor are indicated by x and. , and the windings A and A make up a pair, so as B and B, and C and C,thus forming unit coils respectively. These unit coils are of thewindings A, B and C phases respectively.

In the embodiment in FIG. 22, the flux of one unit magnetic pathintersects windings of a plurality of phases. In other words, fluxes areproduced across the gap of one unit magnetic path by the compositecurrent (composite magneto-motive force) synthesized from currents(magnetomotive forces) of plural phases. Accordingly, the distributionof magnetic fields across the gap, i.e., the direction (rotation angle)of magnetic fields across the gap, is determined directly by the vectorsynthesis of currents of the respective phases. Thus the electricalvector rotation angle directly determines the driving force balancepoint.

This is advantageous over the phase separation type (such as shown inFIG. 21, or the type of tandem connection across phases, or the typehaving magnetic paths respective for phases without connections acrossphases) in which the electrical vector rotation angle θ is oncereflected on the torque produced for each phase and then the drivingforce balance point is determined indirectly as the composite torquebalance. This advantage is enhanced because the construction of theinvention has a plurality of kinds of magnetic teeth for each phase.

FIG. 23 shows a magnetic core pattern for a construction having a largernumber of slots. The number of coincident points M₁ to M_(q) for themagnetic teeth may be arbitrarily determined.

FIG. 24 (a) and (b) show methods of winding applied to the constructionhaving 12 slots. FIG. 24 (a) shows the construction of a 3-phase 2-pole15° pitch winding in which the conductors indicated by alphabets withbars are paired to form unit coils respectively. FIG. 24 (b) shows theconstruction of a 3-phase 4-pole 180° pitch winding. The 2-pole windingapplies to the number of coincident points Q = 2; and the 4-polewinding, to the number of coincident points q = 4. For the constructionhaving a field magnet, the 4-pole winding applies to the number ofcoincident points q = 2. The greater the number of slots, the larger thenumber of unit coils and thus the smaller the distortion of the gapfield distribution, and the better the vector composite effect bycurrents of individual phases synthesized in the direction (rotationangle) of the field across the gap.

FIG. 25 (a), (b) and (c) are diagrams showing the constructions ofinductor type synchronous motors having a DC field magnet, in connectionwith DC field winding methods. FIG. 25 (a) shows the construction of abipolar type having slots 15 for even numbers of field windings 16.Driving winding (armature winding) slots 13₁ to 13_(n) are installedbetween the slots. Often, the driving windings 14₁ to 14_(n) areinserted in the field windings 16 for use in common with the slots. Thedriving windings 14₁ to 14_(n) are wound so that a moving field (ormoving current) is formed in part of the region between field windingslots with the electrical angle 2k₄ (k₄ = 1, 2, 3, ...). The number (q)of tooth coincident points is k₅ k₄ (k₅ ; 1/2 pf the number of fieldslots).

FIG. 25(b) shows the construction of a heteropolar type in which thefield winding slots and the driving winding slots 13₁ to 13_(n) can beused in common. The field windings 16 form magnetic poles N and S withone another. The driving windings 14₁ to 14_(n) are wound in the samemanner as in FIGS. 21 to 24. In this example, the number of windingpoles is twice that in the case of the construction having no DC fieldmagnet. (The number of winding poles does not mean the number of unitmagnetic paths 120, but is the number of magnetic poles produced whencurrent is fed to the winding of one phase in the case of polyphasewinding construction.) For instance, the arrangement shown in FIG. 25(b) is described in relation to the one shown in FIG. 21. The polaritiesof the driving windings a, b and c are inverted from a, b and c. Thatis, the forward polarity symbols a, b and c are given in place of thereverse polarity symbols a, b and c for the driving windings.

FIG. 25(c) shows the construction of a dual pair homopolar typedisclosed in Japan Patent Application No. 48- 31627 (1973); across-sectional view across the plane including the rotation axis isillustrated. The field windings 16a and 16b are of doughnut type (ringtype), and the pattern of inductor cores 100 and 200 is as shown in FIG.23. The field flux passes through a loop: the first inductor 100₂--gap--the second inductor 200₂ --the second inductor 200₁ --the firstinductor 100₁ --the first inductor 100₁ --the yoke 17, and also througha loop: the first inductor 100₂ --the second inductor 200₂ --the secondinductor 200₃ --gap--the first inductor 100₃ --the yoke 17, whereby aradial field of unipolarity is formed across the gap.

Permanent magnets 16a' and 16b' may be installed instead of the fieldwindings 16a and 16b. The driving winding 140 is wound in the form ofpolyphase winding as in FIG. 24(b). The number of polarities of thewindings is made twice the number q of the coincident point M. For thisarrangement, various methods of polyphase winding may be employed suchas in connection with the number of slots made for each pole and eachphase, and short-pitch winding.

In the construction having a DC field magnet, the fluxes φ₁ to φ_(i) aregiven uniformly by DC field means and expressed identically by thefollowing equation:

    φ.sub.i = K.sub.6 I.sub.f a.sub.si (13)

where K₆ : constant, indicating a saturation characteristic dependent onI_(f)

I_(f) : field current

Thus the produced torque is proportional to the driving winding current(when the field current I_(f) is constant), having a proportionalcharacteristic curve as in FIG. 7 even with magnetic saturation takeninto consideration. It is desirable that the DC field is intensified andthe driving winding magneto-motive force is reduced to improve the powerfactor. Therefore the produced torque comes in the range of roportionallinearity. It is most desirable that control accuracy is determined inthe range of good linearity.

Equation (11) is rewritten as follows, where Ψ₀ (I_(f)) exhibits amagnetization characteristic pattern of the straight line C or thesaturation curve B as in FIG. 7.

    τ.sub.dc = A.sub.1 ·Ψ.sub.0 (I.sub.f) ·I.sub.1 (m/2)S.sub.in (θ- θ.sub.m) (14)

where τ_(dc) : torque of DC exciting type motor

A motor of the reluctance type without DC field magnet has aself-exciting characteristic, in which the value I_(E) which correspondsto the excitation component I_(f) is given by the following equation.(Although a strict operational result is supposed to be derived from thepermeance and current distribution of each tooth, such operationalapproach is omitted for the sake of simplicity.)

    I.sub.E = I.sub.1 (m/2)cos (θ- θ.sub.m) (15)

From Eq. (14) the reluctance motor torque τ_(R) is given as

    τ.sub.R = A.sub.1 Ψ.sub.0 (E.sub.E) ·I.sub.1 (m/2)S.sub.in (θ- θ.sub.m) (16) where Ψ.sub.0 (I.sub.E) is a magnetic saturation curve as B in FIG. 7. In the linear proportional region, the torque τ.sub.R is

    τ.sub.R ≈ 2K.sub.R I.sub.1.spsb.2 S.sub.in (θ- θ.sub.m)·cos (θ- θ.sub.m) = K.sub.R I.sub.1.spsb.2 S.sub.in 2(θ- θ.sub.m) (17)

where K_(R) is a constant dependent on the proportional constant m ofA,Ψ₀ (I_(E)).

Here the value 2 (θ- θ_(m)) is the load angle δ_(R) of the reluctancetype motor.

In the magnetic saturation region, the value Ψ₀ (I_(E)) is saturated andthe torque exhibits a current proportional characteristic.

As is apparent from Eqs. (16) and (17), the feed current vector rotationangle θ is accurately reflected on the driving force balance pointθ.sub.τ and the electromagnetic balance oint θ_(m). This principleoriginates from the condition allowing Eqs. (6) and (7) to beestablished, and depends on the accuracy with which whether or not thevector rotation angle can be expressed only by the first order component(ν = 1) and reduced higher order component (ν ≧ 2). The key to thisprinciple lies in Eqs. (4) and (5).

More specifically, it is essential that many kinds of first magneticteeth having many kinds of degrees of coincidence (absolute value andvarying factor and directivity, as well as phase and varying pattern)and having various permeance varying patterns and varying phases, have aflux intersecting relationship with the winding of one phase. Thus"distortivity" of magnetic teeth of a single kind is reduced and theratio of the higher order component (distortion component) to the firstorder varying component (fundamental component) is reduced.Consequently, magnetic circuit distortion is reduced.

For instance, in the method of shifting the tooth phase ζ_(1i), twokinds of teeth are used and the phase difference between the two is setto π/2 whereby the second order component (ν = 2) which is the greatestcause of error is eliminated. In another method, two kinds of teeth areused and the phase difference between the two is set to π/3 whereby thethird order component which is the second greatest cause of error iseliminated. In another method, magnetic teeth of three phases ζ₁₁, ζ₁₂and ζ₁₃ are used to intersect fluxes in numbers W₁, W₂ and W₃ (or toothwidths t₁, t₂ and t₃) whereby a lower order (ν = 2, 3 or ν = 2, 4)component can be markedly reduced.

The higher order component (ν > 2) can be greatly reduced when the phaseζ_(1i) is set to a distribution width ±Δζ = π/3 to π/4, without the needfor eliminating a specific order component. This is available by settingthe coil pitch of the unit coil to an electrical angle of 2π/3 to π (forinstance, one shown in FIGS. 22 and 24, or in FIG. 21 which has a DCfield magnet).

The above can also be realized by the skew of the inductor teeth as inFIG. 19. In the construction in FIG. 16, the winding for one phase ismade to intersect a plurality of unit magnetic paths wherebysubstantially numerous kinds of distributions can be obtained as in FIG.19. Further, higher or lower order components can be markedly reduced byincreasing the number of slots (FIG. 20) and thus increasing the kind ofW_(i).

In another method, the coil pitch is increased to allow unit magneticpaths and gaps to be used in common (polyphase distribution winding)whereby the difference between θ and θ_(m) can be eliminated. This pointcan be absolutely improved by emphasizing polyphase distribution, inaddition to increasing the number of slots. Polyphase distributionwinding (gap-common winding) as in FIGS. 22 and 24 offers a rotatingfield (moving field) where waves of field are uniformly moved orrotated, which rotates or moves in accurate corresondence to the vectorrotation angle of the quantity of feed electricity. The gap field itselfdraws a locus as in FIG. 5.

By the above arrangement, a nondistortion elctromagnetic construction(with higher harmonic components removed, and sinusoidalized internalvelocity electromotive force) can be realized. The key to suchnondistortion electromagnetic construction lies in that the fluxes of aplurality of kinds of magnetic teeth intersect the winding correspondingto one phase. Means to realize this effect include an arrangement forpitches of magnetic teeth, adjustment of tooth widths, skew, increase inthe pitch of unit coils, increase in the number of winding slots,increase in the number of unit coils, and an approach to thedistribution winding.

When power is supplied uniquely and trigonometrically in function to theinductor type synchronous motor of nondistortion construction realizedin the foregoing manner, the electrical vector rotation angle θ of thefeed power AC component becomes accurately coincident with the drivingforce balance point θ.sub.τ. Thus highly accurate resolution control isrealized by a very simple means (a feed power pattern generating means,i.e., a trigonometric function generator and a solid-state switchcircuit operated as a power amplifier means). Furthermore, the currentvalue I (total proportional representative value such as wave heightvalue I_(j)) can be arbitrarily changed or suitably controlled such as,for instance, the current value I is made to correspond to a giventorque or to an accelerating or decelerating speed, or changed at apredetermined time of operation.

I claim:
 1. A fine control system for driving an inductor typesynchronous motor comprising:an inductor type synchronous motor having afirst inductor equipped with driving windings corresponding to m phasesand a group of first magnetic teeth, and a second inductor equipped witha group of second magnetic teeth, m representing a positive integer; aDC power supply; a plurality of switches connected between the DC powersupply and the driving windings for distributing the DC power supplyvoltage to the driving windings of the individual phases; control meansfor generating command pulse trains with meaning of positive andnegative rotation and fine control mode command pulse trains; anddistribution pattern generator means connected to the control means andresponsive to the generating of a fine control mode command pulse trainfor controlling the on-off operation of the pulurality of switches, thedistribution pattern generator means comprising: a carrier pulse traingenerator; gate means connected to the control means and to the carrierpulse train generator and responsive to the generating of a fine controlmode command pulse train for passing a carrier pulse train; a firstpulse train adder connected to the gate means and to the control meansand responsive to the generating of a positive rotation command pulsetrain for summing the positive rotation command pulse rain with acarrier pulse train passed by the gate means; a second pulse train adderconnected to the gate means and the control means and responsive to thegenerating of a negative rotation command pulse train for summing thenegative rotation command pulse train with a carrier pulse train passedby the gate means; a reversible ring counter having a count-up input andcount-down input and connected to the plurality of switches; a firstfrequency divider connected between the first pulse train adder and thecount-up input of the counter; and a second frequency divider connectedbetween the second pulse train adder and the count-down input of thecounter; whereby a step number R in excess of 2m per electrical cyclecan be obtained.
 2. The fine control system recited in claim 1 whereinthe first inductor comprises a group of unit magnetic paths for formingthe driving windings, the unit magnetic paths divided by slots, eachunit magnetic path having a plurality of unit magnetic teeth installedon a surface thereof facing opposite to the group of second magneticteeth, the unit magnetic teeth belonging to the group of first magneticteeth, and the fist inductor further having a group of unit coils, eachof the driving windings of the individual phases being formed of theunit coils in a series-parallel connection.
 3. The fine control systemrecited in claim 2 wherein at least two magnetic teeth of at least oneof the magnetic paths are different in kind from each other.
 4. The finecontrol system recited in claim 2 wherein the group of magnetic teeth ofone phase comprise at least two magnetic teeth different in kind fromeach other and installed on different unit magnetic paths respectively.5. The fine control system recited in claim 1 wherein the pitch of thefirst magnetic teeth is differentiated from that of the second magneticteeth.
 6. The fine control system recited in claim 1 wherein the groupof first magnetic teeth are shifted in sequence in the directionperpendicular to the relative moving direction of the first and secondmagnetic teeth.
 7. The fine control system recited in claim 1 whereinthe group of second magnetic teeth are shifted in sequence in thedirection perpendicular to the relative moving direction of the firstand second magnetic teeth.
 8. The fine control system recited in claim 1wherein at least two of the group of first magnetic teeth have toothwidths different from each other.
 9. The fine control system recited inclaim 1 wherein a DC field means for providing a DC field is disposedbetween the group of first magnetic teeth and the group of secondmagnetic teeth whereby the linearity of the current-torquecharacteristic is improved.